Before discussing the calculation of magnetic components for switching power supplies, let me just quickly go over the basic concepts and definitions. Transformer is a passive device which transfers alternating (AC) electric energy from one circuit into another through electromagnetic induction. It consists of a ferromagnetic core and two or more coils (windings).
A changing current in the primary winding creates an alternating magnetic field in the core. The core multiplies this field and couples most of the flux through the secondary windings. This in turn induces alternating voltage (electromotive force, or emf) in each of the secondary coil according to Faraday's law. Power transformer in SMPS is used to change amplitude of high-frequency pulses by the turns ratio and to provide isolation between circuits.

Note that a transformer can't transfer a DC component of a pulse: in a steady state mode net volt-seconds across any winding should be zero, otherwise the core will soon saturate. DC output voltage can be obtained only by using rectifiers. Nevertheless, an average voltage across a real coil's terminals can be non-zero due to non-zero wire resistance. This DC offset can be used for lossless sensing of an average current across an inductor or a transformer winding with unidirectional current: if you add an RC network parallel to the coil, the voltage across the capacitor will be proportional to the coil's average current. For better thermal stability the wire can be made of low TCR material, such as a copper alloy.


normally involves trade-offs between size, cost and power losses. The main constraint in all cases (except for saturable inductors) is that peak magnetic flux density BMAX should not approach the core material's saturation flux value BSAT. Note that in higher frequencies, core loss rather than saturation can become the main limiting factor for BMAX. The flux change is a function of the applied volt-seconds and the core geometry. Excessive volt-seconds applied to a coil cause core saturation. When it happens, the windings are effectively shorted out.

The table below provides the formulas for BMAX in a steady state operation as function of N×Ac product, applied voltage and frequency for common voltage waveforms. Contrary to popular misconception, BMAX does not depend on the magnetic material properties or air gaps. It does not depend on the transferred power neither. That's why in theory the core size does not depend on the wattage. However, for efficiency and thermal reasons, we have to limit ohmic losses in the wires. That's why magnetics size increases with power. Most textbooks provide formulas for estimation of the core size based on the product of magnetic cross-section area by the window area available for the winding. Unfortunately, this method is not very helpful because these formulas are based on pretty much arbitrary selection of current density and on the assumption of a certain window utilization (fill) factor. In reality, depending on the insulation grade and application requirements, current density in the copper can be selected anywhere between 180 and 500 A/sq.cm. The fill factor likewise can be anywhere between 0.1 and 0.5 depending on the coil construction and insulation. As the result, if you don't have much experience in practical transformer design, you will need to go through several iterations.

Function Waveform BMAX, gauss
Sine wave Sinewave voltage Vrms×108/4.44N×Ac×F
Square wave Square wave voltage Vpk×108/4N×Ac×F
Bipolar pulses with D=Ton/T=Ton×F
PWM voltage Vpk×D×108/2N×Ac×F
Unipolar pulses
with passive reset
Unidirectional voltage pulse Br+Vpk×Ton×108/N×Ac
In these and other equations: V - voltage (volts), N - coil's turns, Ac - core's cross-sectional area (sq.cm), F- frequency (hertz), Br - remanence (gauss)
Here is a quick simplified transformer design procedure:
When you get an actual prototype you can measure the hot spot temperature by placing a thermocouple under the coil and then adjust the design if necessary. S.A.Mulder proposed an empirical formula for thermal resistance of a wire-wound transformer measured at the hot spot: Rth≈(53...61)×Ve-0.54 oC/watt, where Ve- core volume in cubic centimeters [1990 Phillips App Note]. From this, a "rule of thumb" hot-spot temperature rise in Celsius vs. total losses P(watt) is: Trise≈50/sqrt(Ve)×P. Note that the above thermal relationship as well as most textbook procedures related to the magnetics thermal management are applicable to natural convection cooling. For applications with forced airflow or conduction cooling these procedures results in an over-designed part because of an overstated temperature rise.

In general, ideal SMPS transformers need to transfer all energy instantaneously from one winding to another while storing no or little energy in the process. Some topologies do need certain amount of energy stored in magnetizing inductance for a proper operation. Conversely, a power inductor is used in SMPS as an energy storage device. It accumulates energy in the magnetic field as current flows through it, and then transfers it into another circuit during the alternate part of the switching cycle. In power supplies, the inductors are also used for filtering out high frequency currents (in which case they are often called chokes). In power inductors the current rather voltage is controlled. For such "current-driven" coils:
where L - inductance (in henry), Ipk - peak current in amps, B - flux in gauss. All units and formulas in this page are given in CGS (see CGS to SI unit converter). Once you set BMAX and choose a core size, you can find from the above equation the number of turns for the desired inductance L: N=L×Ipk×108/BMAX×Ac.

Note that "L" is not constant. If the current keeps increasing, at some point BMAX will be approaching BSAT and "L" will start dropping. To prevent the magnetic material saturation at a required current, an air gap can be introduced. The length of a net discrete gap: lg≈0.4×π×N×Ipk/BMAX. In practice, the gap may need to be selected slightly larger than the calculated value due to flux fringing. Combining the above equations for N and lg yields: lg≈0.4×π×L×Ipk2/(BMAX×Ac).

The gap is used not only in inductors. It is also often introduced in transformers to increase working flux swing in single-ended topologies, to store more energy in magnetizing inductance, and to stabilize the inductance value. For powder metal materials with a distributed gap and soft saturation curve, the calculation process may take several iterations. In short, you can first pick a powder core based on desired L×Ipk2 by using manufacturer's charts. Then determine the turns Formula for inductor turns, where AL - specific inductance in mH/1000 turns (which is nH/turn) from the data sheet. Then find peak bias H=0.4×π×N×Ipk/le (Oersted), determine the roll-off in percentage of initial permeability, and correct the turns for the desired L.

Below you will find more magnetics theory, transformer and inductor design information, tutorials, tools and free downloads.

Transformer turns and wire calculator
(includes skin effect)

Transformer calculation for various switching regulator topologies

Software to design electrical inductors with powder cores by Magnetics® Inc.

Current transformer design software

Ferrite magnetic calculation tool (includes skin and proximity effects)

Core loss calculator for non-sinusoidal waveforms

(MAG 100A)
Introduction and magnetics basics (design for switching power supplies)

Magnetic core characteristics

Windings data and skin effect

Power supply transformer design

Inductor and flyback transformer design

Magnetic core properties

Eddy current losses in transformer windings

The effect of leakage inductance

Coupled filter inductors

Designing transformers for high frequency dc-dc converters - pdf download

Core selection for flyback and forward converters

SMPS transformer design procedure and equations

Inductor design procedure

Planar power transformers basics and design guide

Electrical transformer: how it works and physical principles